Low Sidelobe Slot Array Antenna Radiating Monopulse Patterns Based on Groove-Transform-Ridge Gap Waveguide

Abstract

In this paper, a W-band unequal power divider is proposed on the basis of groove-to-ridge gap waveguide, and it is employed to design a low sidelobe level (SLL) slot array antenna with monopulse radiation patterns. The proposed unequal power divider is capable of realizing large power ratio and small phase difference. In light of this property, three different unequal power dividers with various output power ratios and phase balance are proposed for designing a low SLL array antenna. Furthermore, a compare network with planar Magic-Tee is designed for realizing the monopulse radiation patterns. Then, a 32 × 32 antenna array consisting of five unconnected layers is designed, fabricated, and measured. The measured results indicate that the relative impedance bandwidth of the antenna is about 5.5%. In addition, the realized gain of the sum pattern at 84 GHz is 35.4 dBi, while the null of the difference pattern is less than 38 dB. Furthermore, the SLL of the antenna array is more than −25 dB on both the E- and H-planes within the operating band.

This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

1. Introduction

In recent years, millimeter wave (mmWave) technology has played an important role in both communication and radar systems [1, 2]. On the one hand, the extremely high data rate transmission in communication systems is indispensable and new requirements are being placed on the integrated and low-cost devices for the applications of satellite traffic [3], ultra-high-definition video [4], and ultrahigh resolution imaging [5]. On the other hand, mmWave technology is required in radar systems to perform efficient detection for long-range unlicensed targets [6]. In these applications, monopulse radiation and low sidelobe level (SLL) performances are essential [7]. Concretely, the monopulse technology is adopted to accurately detect scattered information from targets, and the low SLL technology is the guarantee for low interception and anti-interference performances. Therefore, it is crucial to design an mmWave monopulse antenna with low SLL characteristics.

Compared with the lens and reflector antennas [8], the planar antenna can achieve high gain and monopulse radiation performance while ensuring compact and low-profile structure. By using the microstrip lines or substrate integrated waveguide technologies, the planar array can realize a wideband monopulse radiation performance [9, 10]. Nonetheless, the radiation efficiency of the antenna is reduced because the field leakage and loss in the substrate will be more serious in the millimeter wave band. The novel technology of gap waveguide, which is newly reported in [11, 12], performing enormous advantages such as manufacturing process and low dielectric loss in mmWave band. Thus, several related mmWave slot array antennas were represented in [13–15]. However, these array antennas have a high SLL due to the same power distribution using full-corporate networks.

The nonuniform tapering power distribution is an effective way to suppress the SLL, which is easy in the low frequency band. However, this method is difficult to realize in the mmWave band because of the strict process technology. A common approach for achieving low sidelobe levels (SLL) is the serial-fed slot array [16]. By adjusting each slot’s offset and length based on Taylor synthesis, resonance occurs in each slot, and the amplitude tapering is obtained via varying impedance. However, due to the long-line effect, this method significantly narrows bandwidth. Alternatively, a corporate-feeding network can be employed to achieve array synthesis. While this network can deliver wider bandwidth, it is susceptible to grating lobes due to the quasi-Taylor distribution and makes it difficult to achieve high power ratio dividers and consistent phase. In [17], a trench was introduced into the corporate-feed backed cavity subarray to address E-plane grating lobes. However, this approach only resolves E-plane issues and adds substantial structural complexity.

In [18], the monopulse slot array antenna was rotated to achieve low SLL. Nevertheless, the radiation pattern cannot be improved at every azimuth angle with this method. Another comparable work [19] shows that the low SLL of the box-horn antenna array radiating monopulse beams is obtained by using the hollow-waveguide with adjustable split-block. However, this antenna array is a fully enclosed metallic structure that requires special process to ensure electrical connections between each layer; thus, its manufacturing process is complicated and costly. Recently, a new method has been introduced to reduce the SLL in the mmWave band by only changing the phase from 0 to 180 degrees, which can simplify the design and processing difficulty [20, 21].

In this paper, a novel unequal power divider using groove-to-ridge gap waveguide was systematically introduced, which can realize the large power ration and small phase difference by adjusting the length and height of the ridge. Then, three unequal power dividers with different output power ratios are carefully designed for realizing a W-band low SLL array antenna. Furthermore, a compare network with planar Magic-Tee is designed for a 32 × 32 antenna array consisting of five unconnected layers to obtain the monopulse radiation patterns and low SLL. Simulated and measured results are compared to indicate a good monopulse radiation and low SLL performance.

2. Design of Power Distribution Network

Before designing the unequal power divider and power divider network, we first obtained the dispersion diagrams of gap waveguide unit cell through simulation optimization of Eigenmode solver in CST software, as shown in Figure 1. We noted that the specific analysis and design guidance can be found in literature. It is found that the gap waveguide can achieve a wide stopband from 42 GHz to 123 GHz, although there is a certain gap between the upper and lower layers. At the same time, the gap waveguide allows the transmission of TE₁₀ mode within 54 GHz to 123 GHz, which is also the dominant propagation mode used in the design.

[figure(s) omitted; refer to PDF]

2.1. Design of Unequal Power Divider

As is known to all, the Taylor or Chebyshev distribution is usually utilized to realize the low SLL performance of antenna arrays and the lower SLL required larger power ration. Thus, the key to effectively reduce the SLL of an antenna array is to flexibly control the output power ratio of the unequal power divider and ensure the balance of the output phase and expect to obtain a larger power ration, which requires that the unequal power ratio is capable of control the output power ratio and phase difference.

The designed unequal power divider, shown in Figure 2, employs three different ridges to achieve impedance matching and power distribution. According to the simulated electric field distribution in Figure 3, the varying ridge design leads to an uneven distribution of input power between Port 2 and Port 3. Key parameters influencing the output amplitude and phase differences are analyzed, revealing that the height difference ($∆h=h_{\text{out}1}-h_{\text{out}2}$) contributes to variations in output amplitude, while the length difference ($∆L=L_{\text{out}1}-L_{\text{out}2}$) impacts output phase differences. Therefore, we investigate the reflection coefficient, output amplitude, and phase differences of the divider based on $∆h$ and $∆L$. The remaining parameters in Figure 2 remain fixed and are outlined in Table 1.

[figure(s) omitted; refer to PDF]

Table 1

Geometrical sizes.

Geometrical size	Values (mm)
$W$	0.45
$L_{in}$	1.10
$h_{in}$	0.30
$L_c$	1.28

The influence of parameter $∆h$ on the performance of the proposed unequal power divider is analyzed, as shown in Figure 4. First, the reflection coefficient of the designed unequal power divider with various $∆h$ is shown in Figure 4(a), which shows a good phenomenon that the reflection coefficient has no significant change. Furthermore, the amplitude difference increases observably when the height difference $∆h$ enlarges, as shown in Figure 4(b), which helps achieve a large output power ratio. Concretely, the amplitude difference could be greater than 10 dB when $∆h$ is 0.4 mm. Unfortunately, the output phase difference also increases with the larger $∆h$, as shown in Figure 4(c), resulting in the deterioration of the radiation performance of the array antenna. Thus, it is necessary to compensate the phase difference by adjusting other parameters.

[figure(s) omitted; refer to PDF]

On the other hand, we also study the influence of the length difference $∆L$ on the designed unequal power divider by parameter sweep, as shown in Figure 5. Compared with the height difference $∆h$, the length difference $∆L$ has a more obvious influence on the reflection coefficient, but it is still lower than −10 dB. In addition, the parameter $∆L$ mainly affects the output phase difference, while it is insensitive to the output amplitude difference. Therefore, the output phase difference can be compensated by varying the length difference between the ridges at Port 2 and Port 3.

[figure(s) omitted; refer to PDF]

Based on the analysis of the above two sensitive parameters $∆h$ and $∆L$, we can design an unequal power divider with desired power ratio and phase balance and apply it to the antenna array with low SLL. Usually, when designing the desired power ratio, we first adjust the parameter $∆h$ to obtain the desired output amplitude difference and then change parameter $∆L$ to compensate the output phase difference so that it can reach phase equilibrium (less than $5^{\mathrm{o}}$). In addition, the overall reflection coefficient, the output amplitude and phase differences of the designed unequal power divider are optimized by appropriately adjusting the remaining parameters ($L_{\text{in}}$ and $h_{\text{in}}$).

2.2. Design of the Power Distribution and Comparator Networks

On account of the aforementioned analysis and discussion of unequal power dividers, a 16 × 16 full-corporate-feed distribution network is utilized to construct a 32 × 32 low SLL slot array antenna. The entire distribution network is based on 30-dB Taylor synthesis and is completely composed of the proposed unequal power dividers, as shown in Figure 6. Given that the slot array antennas are symmetric along both the x- and y-axes, only a quarter of the feeding network is shown in Figure 6. The values inside the dashed boxes represent the power ratios required for each corresponding unequal power divider. Despite the need for multiple unequal power dividers in the feed network, only three types are used based on the power distribution criteria to simplify the overall design: 1.5 dB, 3.4 dB, and 4.6 dB. Notably, the reflection coefficients are below −10 dB within the operating band, and the output phase differences are corrected within a 5-degree range.

[figure(s) omitted; refer to PDF]

The comparator network for generating the monopulse radiation pattern is then designed as a critical component for achieving both sum and difference patterns. As shown in Figure 7(a), the network employs a two-layer planar structure, utilizing gap waveguide technology to prevent electrical field leakage between layers. The two-layer planar Magic-Tee is essential for maintaining a low-profile, planar configuration and serves as a critical component for the 3-dB power dividers in the E- and H-planes. As shown in Figure 7(b), it is seen that the path of $P_1^{\prime }⟶P_2^{\prime }$ is constructed in the upper layer, $P_1⟶P_1^{\prime }$ and $P_2^{\prime }⟶P_2$ is realized by means of the slot couplings, and other paths are arranged in the lower layer.

[figure(s) omitted; refer to PDF]

When the sum port is excited, the signals pass through $P_1⟶P_1^{\prime }⟶P_2^{\prime }⟶P_2$ and eventually output from ports ${\Sigma }_1$ and ${\Sigma }_2$. We can find that the output signals have the same amplitudes and inverse phases both in the E- and H-planes. Nonetheless, the inverse phases are compensated because the feed network is mirrored vertically and horizontally, and then a sum beam is generated, as shown in Figure 7(c). On the other hand, when the difference port E is excited, the inverse phase in the H-plane caused by the slot coupling is compensated by the mirror symmetry. Correspondingly, the output phase of the feed network in E-plane is identical; thus, the opposite phase brought by mirror symmetry realizes the difference beam in E-plane, as shown in Figure 7(d). In the same way, the H-plane difference beam will be generated when the difference port H is excited, as shown in Figure 7(e).

3. Design of Antenna Array

3.1. Subarray Element

In our designed antenna array, the basic radiation element is a 2 × 2 slot antenna subarray, as shown in Figure 8. It is found that four slots are excited in one cavity, which has the advantage of inhibiting high grating lobes. We can find that the whole structure consists of three layers, which are the top radiation layer, the backed cavity layer, and the bottom feed network layer. Then, four slots at the top layer are excited with equal amplitude and phase through the coupling hole and the cavity by the gap waveguide. In addition, both horizontal and vertical spacings between the two adjacent slots are set to 2.85 mm, which is less than one wavelength (0.8${\lambda }_0$) for the reason of minimizing the grating lobes. Furthermore, the remaining parameters are listed in Table 2.

[figure(s) omitted; refer to PDF]

Table 2

Dimensions of 2$\times$2 subarray (unit: mm).

$h_t$	$g_t$	$h_B$	$h_p$	$L_{s1}$	$L_{s2}$	$W_{s1}$
1	0.5	0.5	1	2.07	1.81	1
$W_{s2}$	$d_1$	$d_2$	$L_B$	$W_B$	$L_{pin}$	$W_{pin}$
0.4	0.48	1.55	1.61	0.5	1.3	0.45

Then, the subarray element is simulated by the CST Microwave Studio with the periodic boundary condition, and its reflection coefficient and radiation patterns of both E- and H-planes are shown in Figure 9. We can see that the impedance bandwidth of the subarray element covers from 81.4 to 85 GHz. Besides, the designed antenna element has a good radiation performance with a gain of 12.9 dBi at 83 GHz, and the SLL is about −6.8 dB and −13.2 dB for the E- and H-planes, respectively.

[figure(s) omitted; refer to PDF]

3.2. Full Array Configuration

The configuration of the designed low SLL slot array antenna is depicted in Figure 10, which contains five layers including the radiation slot array layer, the cavity layer, two middle power divider network layers, and the bottom sum-difference network. Noted that the radiation slot array is composed of 32$\times$32 slots and the feeding network uses a full-corporate distribution based on the designed unequal power divider. The groove gap waveguide is applied in this antenna array, which leads to a simple assembly with screws and pins instead of welding or any electrical contact between each layer.

[figure(s) omitted; refer to PDF]

4. Experimental Results

The prototype of the low SLL slot array antenna with monopulse radiation patterns is fabricated and assembled by screws, as shown in Figure 11. The fabricated array antenna is made of aluminum, and its effective aperture is 81.6 mm$\times$81.6 mm. Then, the designed array antenna is placed in the Compact Antenna Test Range (CATR) test environment to measure its operating bandwidth and radiation patterns for the sake of testifying the low SLL, high gain, and monopulse radiation performance of the antenna array.

[figure(s) omitted; refer to PDF]

Figure 12 compares the simulated and measured reflection coefficients of the fabricated monopulse antenna, which can be seen that the results for the sum and difference ports are all in good agreement. Concretely, the operating band of the designed antenna at sum and difference pattern covers from 81.5 to 85.8 GHz, corresponding to a fractional bandwidth of 5.1%. Compared with the simulated bandwidth, the measured results are slightly narrower, which is caused by the fabrication and assembly errors.

[figure(s) omitted; refer to PDF]

Then, the sum and difference radiation patterns at three frequencies within the operating band are shown in Figure 13 to demonstrate the monopulse radiation and low SLL features of the designed antenna. We can see that our designed antenna does achieve good monopulse radiation pattern within the operating band, and the null of the difference pattern is below 25 dB than the peak of the sum pattern. On the other hand, the sidelobe level of the E- and H-planes are both below than −25 dB, achieving a low sidelobe level. In addition, the simulated and measured gains of the sum pattern are shown in Figure 14. It can be found that compared with the simulated result, the measured realized gain is reduced by about 1 dB, which is caused by the losses in the processing and also the approach of the amplitude-tapering. Furthermore, the measured aperture efficiency is more than 40% and around 43% for maximum value within the operating band.

[figure(s) omitted; refer to PDF]

Table 3 compares the performance of the proposed antenna with other designs. Notably, this slot array antenna stands out with its exceptionally low SLLs and relatively high efficiency. Moreover, the bandwidth is reasonable and acceptable given the antenna’s low SLL characteristics. The gap waveguide offers significant assembly convenience, as it requires only a few screws rather than diffusion welding or compression fitting for W-band applications.

Table 3

Comparisons of the proposed antenna with different antennas.

References	Freq. (GHz)	Gain (dBi)	SLL (dB)	Beam type	Aperture shape
[13]	60	27	−13.0 (E)−13.0 (H)	Sum	Rect
[18]	73	32.6	−25.4 (E)−13.0 (H)	Sum	Circular
[19]	95	30.5	−20.0 (E)−20.0 (H)	Sumdifference	Rect
[22]	94	35.6	−21.8 (E)−23.4 (H)	Sumdifference	Circular
This work	84	35.4	−25.0 (E)−28.0 (H)	Sumdifference	Rect

5. Conclusion

This paper explores a W-band unequal power divider based on groove-to-ridge gap waveguide, which is employed to design a low SLL slot array antenna producing monopulse patterns. The proposed power divider offers a significant power ratio and minimal phase difference, making it suitable for designing W-band low SLL antennas using gap waveguide technology. Furthermore, a comparator network is developed to achieve monopulse radiation patterns. A five-layer 32 × 32 antenna array is designed and tested, with results demonstrating successful monopulse radiation performance and low SLL.

Acknowledgments

This work was supported by the National Natural Science Foundation of China under Grant numbers 62271050 and 62171027 and Aeronautical Science Foundation under Grant 20200018037002.

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