Control of Current Phase Advancing in Single-Leg Power Converters with the Use of Switched Capacitors

Abstract

The switching frequency is much larger than the frequency of the harmonic voltage source fs≫f , and therefore the current i can be expressed as constant. Because of this, we can use the derived equations also with RLC circuit and a harmonic voltage source. 2.3. [...]the value of the duty cycle which controls the value of the switched capacitor is calculated based on the required capacitance and the values of the capacitors that are switched. [...]we realize that in a real situation is the switched capacitor used with a matrix converter and an active motoric load. [...]the final simulation and experimental verification with dSPACE are done with the matrix converter and induction motor as the active load. See PDF.] Parameters Values Units f 25 Hz R1 59.58 Ω R2 67.38 Ω L1 1210 mH L2 2045 mH C1 2.5 μF C2 200 μF Parameters Values Units Gdc −9.7265×10−5 F Gcϕ −2.4878×105 rad/F Gdϕ 24.1979 rad Gωϕ −0.0571 s Glϕ −2.1227 rad/H Grϕ 0.0580 rad/Ω Gm(dB) L2(mH) −10% 2045 +10% f (Hz) 5 54.1 51 49.3 25 34.6 34.2 33.7 45 34 33.5 33.1 Pm(deg) L2(mH) −10% 2045 +10% f (Hz) 5 87.2 86 85.1 25 63.1 61.9 60.8 45 61.4 60.2 59.1 Parameters Values Units Pn 150 VA Vrms 230 V f 50 Hz Rs 59.58 Ω RS 67.38 Ω Lss 1210 mH LSS 2045 mH Lms 1120 mH LmS 1471 mH J 488×10−6 kg·m2 F 10×10−6 N·m·s pp 1 − Ns/NS 1.6 − Author Contributions Validation, M.P.; formal analysis, R.K. and J.S.; writing-original draft preparation, R.K.; visualization, R.K.; and supervision, B.D. Funding This work was supported by project ITMS: 26210120021, co-funded from EU sources and European Regional Development Fund and projects APVV 0396-15 (Research of perspective high-frequency converter systems with GaN technology) and KEGA 027ŽU-4/2018 (Modelling, Design and Implementation of the Modern Method in the Educational Process of the Technical Faculties Focusing on Discrete Control of Power Systems).

1. Introduction

A topology of the single-leg matrix converter (MxC) is derived from a single-leg VSI converter system with input rectifier [1,2,3]. The analysis of such a MxC topology scheme has been done [4,5]. The advantages of those schematics are presented by the absence of the input rectifier and bulky filtering capacitor and consequently higher efficiency. This is possible by switching of both reversible connected switches (metal-oxide-semiconductor field-effect transistors ), [6]. Using switched capacitors for creating of quadrature supply or two/phase induction motor fed by VSI converter is described in [7]. For IM fed by single-leg MxC, please refer to the following works [8,9].

The use of switched capacitors has been previously published for VSI supply systems in [10,11,12], for single-leg VSI in [7], and for single-phase VSI in [7,11]. All these applications with switched capacitors are derived for steady state with real-time closed-loop control. The use of switched capacitor control for single-leg VSI has not yet been published. For a single-leg matrix converter, the use of switched capacitors is described in the authors’ recent work [8,9].

These MxC or VSI converters require controlling the capacitance due to the single-leg output of the converter when using two-phase passive or active loads. The auxiliary phase is connected in series with switched capacitors to the converter output in comparison to the main phase which is connected directly to converter output. Multiple topologies of switched capacitors exist with different configurations of switches and capacitors with different properties.

This paper deals with phase shift control of current in a single-leg matrix converter with a two-phase output. A known topology of two parallel connected capacitors and switches is used. Two different control methods are used for the current phase advancing. The first method uses open-loop control, where the required 90 degrees phase shift between the current in the main and auxiliary phase is calculated based on the parameters of the load (resistance and inductance). A novel method for control of the desired variable switched capacitance is also derived for the use in the open-loop control. The closed-loop control of current phase shift is inspired by the phase-locked loop.

2. Principle of Operation

The switched capacitors are comprised of one large capacitor and one small capacitor and two switches (Figure 1a). During the operation, only one switch is in the on-state at a time. A single-pole-double-throw switch is used in our analysis and modeling of the switched capacitors (Figure 1b).

The switch needs to be able to block the voltage of both polarities and conduct the current in both directions. One of this type of switch, two anti-series emitter connected insulated-gate bipolar transistors (IGBT), with the equivalent symbol used in this paper, is shown in Figure 2.

2.1. Method of Conservation of Energy

From the potential energy stored in the capacitors can be deduced the equation for the controlled switched capacitance [7]. The capacitors potential energy is

E=12C^v2,

where E is the energy stored in the capacitor, C is the capacitance, and v the voltage of the capacitor. The schematic in Figure 3a shows the voltage source connected with the capacitor.

The average voltages on the capacitors are

_vC1=DV,_vC2=(1−D)V,

where D is the duty cycle, V is the constant voltage on the voltage source, and ⟨_vC1⟩ and ⟨_vC2⟩ are the average voltages on the capacitors.

Then, the total energy stored in the two capacitors is

E=_E1+_E2.

Combining Equations (1)–(3) yields the value of the switched capacitance as

_CSW=_C1 ^D2+_C2(^1−D)2.

The equivalent switched capacitance _CSW with voltage source is shown in Figure 3b. The capacitance _CSW needs to equal C.

From Equation (4), the value of duty cycle can be derived as

D=_C2±_CSWC1+_C2−_{C1 C2C1}+_C2.

2.2. Averaging Method

We now use a current source as shown in Figure 4 [9,13]. The voltage on capacitor is given as

_vCt=1C∫itdt,

where i(t) is the current in the capacitor C. We can consider the capacitor current as constant values at sufficient high switching frequency i(t)=I .

In the first interval, as shown in Figure 4b, the average current flowing in the capacitor _C1 is given as

⟨_iC1⟩=DI,

where I is the value of a constant current source. During the second interval, the average current on the capacitor _C2 is given as

⟨_iC2⟩=(1−D)I.

We can now derive the voltages on the capacitors _C1 and _C2 from Equations (6)–(8) as

_vC1t=1_C1DIt,

_vC2t=1_C21−DIt,

where t is time. We can look at the capacitor voltages as constant values at sufficient high switching frequency _vC1t=_VC1 , _vC2t=_VC2 .

The average voltage during one switching period from capacitor _C1 is

_vC1=D_VC1,

and from the capacitor _C2

_vC2=(1−D)_VC2.

The average voltage on the switched capacitor is then

_vCsw=D_VC1+(1−D)_VC2.

The voltage on the single capacitor C needs to equal the voltage on the switched capacitor _CSW

_vCt=_vCswt,

which is equal to the average voltage

_vCt=_vCsw.

Voltage on the capacitor C is given as

_vCt=1CIt.

Substituting Equations (9), (10), (13) and (16) into Equation (15) yields

1_CSW=1_C1^D2+1_C2^(1−D)2

The duty cycle can be calculated from Equation (17) as

D=_C1+_C1C224^_C22−4_C1+4_C21_C2−1_CSWC1+_C2.

Let us now look at Figure 5, where we have an RL load connected in series with the capacitor and a harmonic voltage source. The differential equation, describing the RLC circuit in Figure 5a, is

vt=Rit+Lditdt+_vCt,

and in the integrodifferential form as

vt=Rit+Lditdt+1C∫itdt.

A similar equation applies for the circuit with the switched capacitor in Figure 5b.

vt=Rit+Lditdt+_vCsw,

where the voltage _vC(t) equals ⟨_vCsw⟩ , when we neglect the switching ripple. The capacitors _C1 and _C2 voltages are substituted into Equation (21) as

vt=Rit+Lditdt+D_vC1t+1−D_vC2t.

Using Equations (9) and (10), we get the integrodifferential equation as

vt=Rit+Lditdt+^D2 _C1∫itdt+^(1−D)2 _C2∫itdt=Rit+Lditdt+^D2 _C1+^(1−D)2 _C2∫itdt.

Substituting Equation (17) into Equation (23) yields

vt=Rit+Lditdt+1_CSW∫itdt.

Equation (24) (Figure 5b) with switched capacitors is equivalent to Equation (20) (Figure 5a) with a single capacitor. The average value of the capacitance _CSW in the switching capacitors is the same as the value of capacitance C in a single capacitor.

The current i stays almost constant thanks to the inductance L for one switching period. The switching frequency is much larger than the frequency of the harmonic voltage source _fs≫f , and therefore the current i can be expressed as constant. Because of this, we can use the derived equations also with RLC circuit and a harmonic voltage source.

2.3. Comparison of the Methods

The circuit in Figure 6 was simulated in MATLAB/Simulink (R2018a, MathWorks, Natick, MA, USA) environment with the following parameters: _R1=70.53Ω,_R2=52.9Ω,_L1=170mH,_L1=120mH,_C1=5μF and _C2=220μF. The calculated value for the capacitor is _CSW=33.867μF for the frequency f=40Hz .

The first simulation results, shown in Figure 7, were gained using the method of conservation of energy. The simulation shown in Figure 8 was conducted using the averaging method. The calculation of the capacitance needed for the phase shift is described in the next section. The variables in the figures are as follows: _imain is the current in the main phase, _iaux is the current in the auxiliary phase with a normal capacitor (Figure 5a) and _iauxsw is the current in auxiliary phase with switched capacitors (Figure 5b).

As we can see, the first method does not have an exact phase shift of 90 degrees between the two currents. On the other hand, the second method has an exact phase shift of 90 degrees.

The current in auxiliary phase with a single capacitor matches the current with switched capacitors.

The difference in the value of the set (calculated) capacitance can be seen in Figure 9. The current i does not charge the capacitors _C1 and _C2 to the same voltage value and this is the reason the first method fails, as it assumes the same voltage on both capacitors.

3. Control of Phase Advancing with Switched Capacitors

The equations for the switched capacitors are now used to control the phase shift of the current. Figure 6 shows a harmonic voltage source with two-phase RL load (main phase _{R1 L1} and auxiliary phase _{R2 L2} ) and a switched capacitor connected in series with the auxiliary phase. During the change of frequency, the phase shift is always constant at 90 degrees. The following equations describe the calculation of the capacitance

α=arctanω_L1R1,

where ω is the angular frequency.

_Zaux=_R2sinα,

where _Zaux is the impedance of the auxiliary phase.

The value of the capacitance is then calculated as

C=1ω(_Zauxcosα+ω_L2).

The calculation of the capacitance for 90 degrees phase shift is also shown in the vector diagram in Figure 10.

For a variable phase shift, as shown in Figure 11, the following equation can be used for the calculation of capacitance

β=φ−α,

γ={π2−β,β>0β,β<0,

C={1ω(_Zauxcosγ+ω_L2),β>01ω(_Zauxsinγ+ω_L2),β<01^ω2 _L2,β=0.

3.1. Open Loop Control

In open-loop control, the 90-degree phase shift is calculated from precise values of the system parameters (frequency, inductance, and resistance) [9]. We can see the open-loop scheme and simplified control diagram in Figure 12 and Figure 13. First, the value of the capacitance is calculated based on these parameters. Second, the value of the duty cycle which controls the value of the switched capacitor is calculated based on the required capacitance and the values of the capacitors that are switched. These methods were described in previous chapters. The advantage is that there are no stability issues as in a feedback loop, and also no dynamics response of the control output (duty cycle). The disadvantage is that we need to know the parameters of the system, and, if the parameters of the system (inductance and resistance) change over time, the phase shift will also change from the set value of 90 degrees. The method of open-loop control is used in Section 2.3.

3.2. Closed-Loop Control

The closed-loop (feedback) control [14] of the phase shift was inspired by an analog phase-locked loop PLL [15]. We can see the complete control scheme with the circuit in Figure 14. The simplified block diagram is shown in Figure 15. Similar to a PPL, we also require a phase detector to compare the phase shift between main and auxiliary phase currents. A simple mixer (multiplication) is used as the phase detector to compare the phase signals. The current signals in the main and auxiliary phase have different magnitudes. This, however, would also affect the phase detector output, and we want to compare only the phase shift. To solve this problem, two PLLs with automatic gain control AGC are used for measurement of the current signals. The PLLs have unity magnitude outputs thanks to AGC. Another benefit of the PLLs is the removal of noise and current ripple from the signal.

3.2.1. Linearization of the Phase Detector

The large signal phase detector output in Figure 15 can be written as follows

_ϕe=sin(ωt+_ϕmain)cos(ωt+_ϕaux−^φ*)=sin(ωt+_ϕmain)cosωt+_ϕaux−π2=sin(ωt+_ϕmain)sin(ωt+_ϕaux),

where t is time, ω is angular frequency, _ϕmain is the current phase shift in the main phase, _ϕaux is the current phase shift in auxiliary phase, and ^φ*=π/2 is the desired phase shift difference between phases.

The phase shift variables can be written as

_ϕmain=_ϕ¯main+_ϕ˜main,_ϕaux=_ϕ¯aux+_ϕ˜aux,_ϕe=_ϕ¯e+_ϕ˜e,

where _ϕ¯main , _ϕ¯aux , and _ϕ¯e , are the large signal components at the operating point and _ϕ˜main , _ϕ˜aux , and _ϕ˜e , are the small signal components (perturbations) around the operating point at equilibrium.

We can substitute the variables from Equarion (32) into Equation (31) of the phase detector. For simplification, we assume in our derivation of the linear phase detector that the large signal phase shift in the main phase is _ϕ¯aux=0 and in the auxiliary phase _ϕ¯aux=π/2 at equilibrium, then

_ϕe=sin(ωt+_ϕ¯main+_ϕ˜main)sin(ωt+_ϕ¯aux+_ϕ˜aux)=sin(ωt+_ϕ˜main)sinωt+π2+_ϕ˜aux.

The solution of Equation (33) yields

_ϕe=12sin(2ωt+_ϕ˜main+_ϕ˜aux)+sin(_ϕ˜main−_ϕ˜aux).

As we can see, our solution has only small signal components, but also two nonlinearities.

The first nonlinearity with double frequency ac component

sin(2ωt+_ϕ˜main),

can be removed with a low-pass loop filter.

The second nonlinearity can be removed if we assume that the small-signal phase shift difference is sufficiently small _ϕ˜main−_ϕ˜aux≈0 , then we can write

sin(_ϕ˜main−_ϕ˜aux)≈_ϕ˜main−_ϕ˜aux.

Then, in the steady state (equilibrium), is the small signal output (error signal) around the operation point of the phase detector equal to

_ϕ˜e=_KPD(_ϕ˜main−_ϕ˜aux)=12(_ϕ˜main−_ϕ˜aux),

where _KPD is the phase detector gain.

3.2.2. Linearization of the Controlled System

To be able to use classical control system theory, we need to have a linear dynamic system. One way to achieve this is to use linearization [16]. We use Taylor series expansion around a specific quiescent operating point. Only the first order term in the series can be used because of the linearity requirement, i.e. higher order terms represent nonlinearities.

The equation for the switching capacitor is derived in the previous section (Equation (17)). The phase shift of the auxiliary phase current is calculated as

_ϕaux=−arctanω_L2−1ωC_R2

As we can see, the phase shift is a function of the angular frequency ω , capacitance C, inductance _L2 , and resistance _R2 . We control the phase shift with the change in the capacitance C which is again controlled by the duty cycle D. The changes in angular frequency ω , inductance _L2 , and resistance _R2 represent disturbances in our control system output which is the phase shift _ϕaux .

We linearize about one variable in the case of the switching capacitor

D=D¯+d˜,

and about four variables in the case of the phase shift of the auxiliary phase

ω=ω¯+ω˜,C=C¯+c˜,_L2=_L¯2+_l˜2,_R2=_R¯2+_r˜2,

where D¯ , ω¯ , C¯ , _L¯2 , and _R¯2 are the equilibrium operating point variables and d˜ , ω˜ , c˜ , _l˜2 , and _r˜2 are the small signal perturbation variables around the operating point.

The first order Taylor series approximation for the equation of the switching capacitance (Equation (17)) is

C=C(D¯)+_{∂C(D)∂DD¯}·d˜+higherorderterms(nonlinearities),

where the equilibrium operating point of the calculated switching capacitance is

C(D¯)=_{C1 C2C2} ^D¯2+_C1 ^(1−D¯)2

and the slope (tangent line) for the linear small signal value of c˜ , which is dependent on the small change of the duty cycle d˜ , is

_{∂C(D)∂DD¯}=−2_{C1 C2}(_C2D¯−_C1(1−D¯))^{(_C1(1−D¯)+_C2D¯)2}.

Again, we use Taylor series and linearize the equation of phase shift (37) about four variables and neglect the higher order terms.

_ϕaux=_ϕaux(ω¯,C¯,_L¯2,_R¯2)+_{∂ϕaux(ω,C,L2,R2)∂Cω¯,C¯,L¯2,R¯2}·c˜+_{∂ϕaux(ω,C,L2,R2)∂ωω¯,C¯,L¯2,R¯2}·ω˜+_{∂ϕaux(ω,C,L2,R2)∂L2ω¯,C¯,L¯2,R¯2}·_l˜2+_{∂ϕaux(ω,C,L2,R2)∂R2ω¯,C¯,L¯2,R¯2}·_r˜2+higherorderterms(nonlinearities),

where the equilibrium operating point of the phase shift is

_ϕaux(ω¯,C¯,_L¯2,_R¯2)=−arctanω¯_L¯2−1ω¯C¯_R¯2.

The small signal phase shift is a function of the linear combination of small signal changes in capacitance c˜ , angular frequency ω˜ , inductance _l˜2 , and resistance _r˜2 . These variables represent four independent inputs.

The slope of the contribution of capacitance c˜ to phase shift is

_{∂ϕaux(ω,C,L2,R2)∂Cω¯,C¯,L¯2,R¯2}=−1ω¯_R¯2 ^{C¯2ω¯_L¯2R¯2−1ω¯_{R¯2 C¯2}2}+1.

The slope of the contribution of angular frequency ω˜ to phase shift is

_{∂ϕaux(ω,C,L2,R2)∂ωω¯,C¯,L¯2,R¯2}=_{L¯2 R¯2}+1^ω¯2 _{R¯2 C¯2}−1^{ω¯_L¯2R¯2−1ω¯_{R¯2 C¯2}2}+1.

The slope of the contribution of inductance _l˜2 to phase shift is

_{∂ϕaux(ω,C,L2,R2)∂L2ω¯,C¯,L¯2,R¯2}=−ω¯_R¯2^{ω¯_L¯2R¯2−1ω¯_{R¯2 C¯2}2}+1.

The slope of the contribution of resistance _r˜2 to phase shift is

_{∂ϕaux(ω,C,L2,R2)∂R2ω¯,C¯,L¯2,R¯2}=1ω¯_{R¯22 C¯2}−ω¯_L¯2R¯22−1^{ω¯_L¯2R¯2−1ω¯_{R¯2 C¯2}2}+1.

From these equations can be derived the transfer functions for control and the disturbances.

Phase shift capacitance to output transfer function:

_Gcϕ(s)=_{ϕ˜(s)c˜(s)ω˜=0l˜2=0r˜2=0}=_{∂ϕaux(ω,C,L2,R2)∂Cω¯,C¯,L¯2,R¯2}.

Phase shift duty cycle to capacitance transfer function:

_Gdc(s)=c˜(s)d˜(s)=−2_{C1 C2}(_C2D¯−_C1(1−D¯))^{(_C1(1−D¯)+_C2D¯)2}.

Phase shift control to output transfer function:

_Gdϕ(s)=ϕ˜(s)d˜(s)=_Gdc(s)_Gcϕ(s).

Phase shift angular frequency to output transfer function:

_Gωϕ(s)=_{ϕ˜(s)ω˜(s)c˜=0l˜2=0r˜2=0}=_{∂ϕaux(ω,C,L2,R2)∂ωω¯,C¯,L¯2,R¯2}.

Phase shift inductance to output transfer function:

_Glϕ(s)=_{ϕ˜(s)l2˜(s)ω˜=0c˜=0r˜2=0}=_{∂ϕaux(ω,C,L2,R2)∂L2ω¯,C¯,L¯2,R¯2}.

Phase shift resistance to output transfer function:

_Grϕ(s)=_{ϕ˜(s)r2˜(s)ω˜=0l˜2=0c˜=0}=_{∂ϕaux(ω,C,L2,R2)∂R2ω¯,C¯,L¯2,R¯2}.

The small signal variables ω˜(s) , _l2˜(s) , and _r2˜(s) represent disturbances of the controlled system and propagate to the output _ϕ˜aux(s) through the transfer function (gain) of _Gωϕ(s) , _Glϕ(s) , and _Grϕ(s) .

The phase shift variation _ϕ˜aux(s) can be expressed as a linear combination of four independent inputs

_ϕ˜aux(s)=_Gcϕ(s)c˜(s)+_Gωϕω˜(s)+_Glϕ(s)_l2˜(s)+_Grϕ(s)_r2˜(s)=_Gcϕ(s)_Gdc(s)d˜(s)+_Gωϕω˜(s)+_Glϕ(s)_l2˜(s)+_Grϕ(s)_r2˜(s).

3.2.3. Control Loop Design

The gain of the phase detector is

_GPD(s)=12.

The designed compensator consists of a low pass filter, which is used for removal of the ac component (second harmonics) from the phase detector output (error signal) and a PI controller.

The transfer function of the second order low pass filter:

GF(s)=KF τFs+12,

where _KF=2 is the filter gain and _τF=0.1s is the filter time constant.

The transfer function of the used PLLs is

_GPLL(s)=_ϕout(s)_ϕin(s)=_KPs+_KI^s2+_KPs+_KI,

where _KP=166.6 and _KI=27,755.55 are the coefficients of the function.

The transfer function of PI controller is

_GPI(s)=_KP+_KI1s,

where _KP=0.05 is the proportional coefficient, and _KI=0.35 is the integral coefficient.

The controller and the whole compensation network are designed with sufficient amplitude and phase margin for robustness of the control feedback loop in the case of changes in the parameters of the system and good rejection of the disturbances.

The open-loop gain transfer function is

_Go(s)=_GPLL(s)_GPD(s)_Gc(s)_Gdϕ(s)=_GPLL(s)_GPD(s)_GF(s)_GPI(s)_Gdc(s)_Gcϕ(s)

The closed-loop control transfer function is

G(s)=_ϕ˜aux(s)_ϕ˜main(s)=_Gdϕ(s)1+_Go(s)=_Gdc(s)_Gcϕ(s)1+_Go(s).

The negative feedback reduces the influence of disturbances in the controlled system.

The closed-loop transfer function of angular speed disturbance rejection is

_Gω(s)=_ϕ˜aux(s)ω˜(s)=_Gωϕ1+_Go(s).

The closed-loop transfer function of inductance disturbance rejection is

_Gl(s)=_ϕ˜aux(s)_l˜2(s)=_Glϕ1+_Go(s).

The closed-loop transfer function of resistance disturbance rejection is

_Gr(s)=_ϕ˜aux(s)_r˜2(s)=_Grϕ1+_Go(s).

The sensitivity transfer function is

_Gs(s)=11+_Go(s).

The phase shift variation _ϕ˜aux(s) with negative feedback and all disturbances is then

_ϕ˜aux(s)=G(s)_ϕ˜main(s)+_Gω(s)ω˜(s)+_Gl(s)_l˜2(s)+_Gr(s)_r˜2(s)=_Gdc(s)_Gcϕ(s)1+_Go(s)_ϕ˜main(s)+_Gωϕ1+_Go(s)ω˜(s)+_Glϕ(s)1+_Go(s)_l˜2(s)+_Grϕ(s)1+_Go(s)_r˜2(s)

The closed-loop performance of the system can be seen from the Bode plot of the open-loop gain _Go , shown in Figure 16, and from the step response G of feedback systems, shown in Figure 17. The rejection of the feedback system to disturbances can be seen in the step response of the sensitivity transfer function _Gs .

A linearized block diagram of the feedback system is shown in Figure 18 and parameters of the system are shown in Table 1. The calculated gain functions for the operating point are shown in Table 2.

The robustness of the designed PI controller and the whole feedback loop to changes of the system parameters can be seen on the Gain margins Gm and phase margins Pm , in Table 3 and Table 4.

4. Simulation of the System with an Active IM Load

Simulation of single-leg two-phase matrix converter has been done with an active two-phase induction motor load using MATLAB/Simulink environment. The principal scheme is shown in Figure 19 [1,2,3,17,18,19,20]. We have two feedback loops. The first feedback loop controls the common current of both phases with a hysteresis controller. The second control loop controls the phase shift of the currents in both phases. The control methods are described in previous sections. The simulation results are shown in Figure 20, Figure 21 and Figure 22. The calculation time step of the simulation is _Ts=1×¹⁰⁻⁵s .

The parameters for the two-phase induction motor used in simulation and experimental verification are shown in Table 5.

5. Experimental Verification and Measurement

Experimental verification was done with the help of dSPACE hardware interface. The connection with a physical matrix converter and induction motor can be seen in the block scheme in Figure 23. Hall effect current probes with amplifier where used for current measurement and differential voltage probes for voltage measurement. The current and voltage probes were connected to the analog input of dSPACE hardware. The digital outputs of dSPACE were used as the control outputs for the switched capacitors current phase control and hysteresis current control and connected to the transistor drivers. Emitter connected IGBT transistors were used as bidirectional switches for the single-leg matrix converter output and for the switched capacitors. The complete real-time control and measurement were done thru the dSPACE hardware and MATLAB/Simulink, ControlDesk software. The control algorithms were set up in the MATLAB/Simulink environment, and after compilation used in dSPACE. The measured data were saved and then plotted in MATLAB. The calculation time step of dSPACE was _Ts=2×¹⁰⁻⁵s . The calculation time step affects the hysteresis band, that is the reason that the current ripple is higher in real measurement than in the simulation. The experimental results are shown in Figure 24, Figure 25 and Figure 26.

6. Conclusions

This paper brings analysis, modeling, computer simulation, and experimental verification of an enhanced one-leg two-phase matrix converter by a switched capacitor supplied by a symmetrical two-phase network with a neutral point. One of the main contributions is the combined control of the auxiliary phase advancing with 90 degrees phase shift under the entire range of the load operation. Against works with single-leg VSI converters (e.g., [1,2,3]), this paper also completes the topology by switched capacitors controlled to obtain quadrature current for the two-phase output.

Two new methods for controlling the phase shift with the auxiliary phase capacitor have been developed. The first method uses the open-loop control where the phase shift is controlled with a calculated value of capacitance for known parameters of the load (resistance and inductance) and the frequency. The value of capacitance is then set in the switched capacitor. The second method uses a negative feedback loop, where the phase shift automatically tunes to 90 degrees. Improved design of the switching capacitor has also been done with the use of the averaging method to get precise values of the switched capacitance.

Regarding electromagnetic interfacing and compatibility, the solving of those is a demanding problematic [21,22], and was not the subject of this article. Supposed applications of this investigated type of MxC converter are in the field of supply of two-phase induction motors for, e.g., pumps, compressors, etc. in a resident environment. A measurement of electromagnetic interference (EMI)/electromagnetic compatibility (EMC) is important for the prototype version of the converter. We are preparing the infrastructure for such a measurement on real prototype completed by EMI filter at the EMI/EMC chamber in the near future.

The simulations have been done in the MATLAB/Simulink environment. Due to simplification, a passive RL load and harmonic voltage source were first used for the analysis, modeling, and simulation with the switched capacitor. However, we realize that in a real situation is the switched capacitor used with a matrix converter and an active motoric load. Therefore, the final simulation and experimental verification with dSPACE are done with the matrix converter and induction motor as the active load.

The derived control methods with the switched capacitor can also be used with a different type of power converter such as the single-leg two-phase voltage source inverter.

References

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Parameters	Values	Units
f	25	Hz
R1	59.58	Ω
R2	67.38	Ω
L1	1210	mH
L2	2045	mH
C1	2.5	μF
C2	200	μF

Parameters	Values	Units
Gdc	−9.7265×10−5	F
Gcϕ	−2.4878×105	rad/F
Gdϕ	24.1979	rad
Gωϕ	−0.0571	s
Glϕ	−2.1227	rad/H
Grϕ	0.0580	rad/Ω

Gm(dB)		L2(mH)
		−10%	2045	+10%
f (Hz)	5	54.1	51	49.3
	25	34.6	34.2	33.7
	45	34	33.5	33.1

Pm(deg)		L2(mH)
		−10%	2045	+10%
f (Hz)	5	87.2	86	85.1
	25	63.1	61.9	60.8
	45	61.4	60.2	59.1

Parameters	Values	Units
Pn	150	VA
Vrms	230	V
f	50	Hz
Rs	59.58	Ω
RS	67.38	Ω
Lss	1210	mH
LSS	2045	mH
Lms	1120	mH
LmS	1471	mH
J	488×10−6	kg·m2
F	10×10−6	N·m·s
pp	1	−
Ns/NS	1.6	−

Author Contributions

Validation, M.P.; formal analysis, R.K. and J.S.; writing-original draft preparation, R.K.; visualization, R.K.; and supervision, B.D.

Funding

This work was supported by project ITMS: 26210120021, co-funded from EU sources and European Regional Development Fund and projects APVV 0396-15 (Research of perspective high-frequency converter systems with GaN technology) and KEGA 027ŽU-4/2018 (Modelling, Design and Implementation of the Modern Method in the Educational Process of the Technical Faculties Focusing on Discrete Control of Power Systems).

Conflicts of Interest

The authors declare no conflict of interest.

© 2018 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open access article distributed under the terms and conditions of the Creative Commons Attribution (CC BY) license (http://creativecommons.org/licenses/by/4.0/).

References

1. Chomat, M.; Lipo, T. Adjustable-speed single-phase IM drive with reduced number of switches. In Proceedings of the 2001 IEEE Industry Applications Conference, Chicago, IL, USA, 30 September-4 October 2001; pp. 1800-1806.

2. Chomat, M.; Lipo, T. Adjustable-speed drive with single-phase induction machine for HVAC applications. In Proceedings of the 2001 IEEE 32nd Annual Power Electronics Specialists Conference, Vancouver, BC, Canada, 17-21 June 2001; pp. 1446-1451.

3. Chomat, M.; Lipo, T. Adjustable-speed single-phase IM drive with reduced number of switches. IEEE Trans. Ind. Appl. 2003, 39, 819-825.

4. Dobrucky, B.; Laskody, T.; Prazenica, M. A novel supply system for two-phase induction motor by single leg matrix converter. Elektron. Elektrotech. 2015, 21, 13-16.

5. Stefanec, P.; Dobrucky, B. One leg MxC analysis and modelling. In Proceedings of the 2015 International Conference on Electrical Drives and Power Electronics (EDPE), Tatranska Lomnica, Slovakia, 21-23 September 2015.

6. Kascak, S.; Dobrucky, B.; Prazenica, M.; Radvan, R.; Spanik, P. Bidirectional Switch with the Use of Reverse Function Mode in MOSFET Transistors; Utility Model No. 6899; Industrial Property Office of the Slovak Republic: Banská Bystrica, Slovakia, 2014; (In Slovak Language).

7. Vodovozov, V.; Lillo, N.; Raud, Z. Single-phase electric drive for automotive applications. In Proceedings of the 2014 International Symposium on Power Electronics, Electrical Drives, Automation and Motion SPEEDAM, Ischia, Italy, 18-20 June 2014.

8. Konarik, R. Single-Leg Matrix Converter: Analysis, Modeling, and Feedback Control. Ph.D. Thesis, Department of Mechatronics and Electronics, Faculty of Electrical Engineering, University of Zilina, Žilina, Slovakia, 2018.

9. Konarik, R.; Dobrucky, B.; Prazenica, M.; Kascak, S. Improved design of switched capacitors for power electronics converters. In Proceedings of the 2018 ELEKTRO, Mikulov, Czech Republic, 21-23 May 2018; pp. 1-6.

10. Lettenmaier, T.; Novotny, D.; Lipo, T. Single phase induction motor with an electronically controlled capacitor. In Proceedings of the Conference Record of the 1988 IEEE Industry Applications Society Annual Meeting, Pittsburgh, PA, USA, 2-7 October 1988.

11. Suciu, C.; Kansara, M.; Holmes, P.; Szabo, W. Performance enhancement of an induction motor by secondary impedance control. IEEE Trans. Energy Convers. 2002, 17, 211-216.

12. Danila, A.; Margineanu, I.; Campeanu, R.; Suciu, C.; Boian, I. The optimization of the single/two phase induction motor start-up with electronically switched capacitor. In Proceedings of the 2008 IEEE International Conference on Automation, Quality and Testing, Robotics, Cluj-Napoca, Romania, 22-25 May 2008.

13. Suciu, C.; Kansara, M.; Holmes, P.; Szabo, W. Phase advancing for current in R-L circuits using switched capacitors. Electron. Lett. 1999, 35, 1296.

14. Franklin, G.F.; Powell, J.D.; Emami-Naeini, A. Feedback Control of Dynamic Systems, 7th ed.; Pearson: London, UK, 2014.

15. Gray, P.; Hurst, P.; Lewis, S.; Meyer, R. Analysis and Design of Analog Integrated Circuits, 5th ed.; Wiley: Hoboken, NJ, USA, 2009.

16. Erickson, R.W.; Maksimovic, D. Fundamentals of Power Electronics, 2nd ed.; Springer: Berlin, Germany, 2001.

17. Zuckerberger, A.; Weinstock, D.; Alexandrovitz, A. Single-phase matrix converter. IEE Proc. Electr. Power Appl. 1997, 144, 235-240.

18. Reveles-Miranda, M.; Flota-Bañuelos, M.; Chan-Puc, F.; Pacheco-Catalán, D. Experimental Evaluation of a Switching Matrix Applied in a Bank of Supercapacitors. Energies 2017, 10, 2077.

19. Peretti, G.; Romero, E.; Marqués, C. On the ability of oscillation-based test for detecting deviation faults in switched-capacitor ladder filters. Electr. Eng. 2007, 90, 127-141.

20. Rajki, I.; Vas, P. Increasing the starting torque of capacitor-run split-phase induction machines by a positive temperature coefficient (PTC) resistor. Arch. Elektrotech. 1984, 67, 65-69.

21. Tarateeraseth, V.; See, K.Y.; Canavero, F.G.; Chang, R.W.Y. Systematic Electromagnetic Interference Filter Design Based on Information From In-Circuit Impedance Measurements. IEEE Trans. Electromagn. Compat. 2010, 52, 588-598. [Green Version]

22. Saponara, S.; Ciarpi, G.; Groza, V.Z. Design and experimental measurement of EMI reduction techniques for integrated switching DC/DC converters. Can. J. Electr. Comput. Eng. 2017, 40, 116-127.